Linear amplifier with distortion correction

ABSTRACT

An a-c linear amplifier with two push-pull-connected output transistors includes an ancillary transistor which, during saturation of one of the output transistors and the simultaneous cutoff of a controlled transistor driving these output transistors through a set of pilot transistors, conducts to maintain the base potential of this controlled transistor at its conduction threshold for preventing a distortion of the end of the flattened peak of the corresponding half-cycle of the output voltage. The ancillary transistor has one of its input leads maintained at a fixed biasing potential and the other of its input leads connected to a point of output-dependent variable voltage that allows its conduction during the flattened peak only.

FIELD OF THE INVENTION

My present invention relates to an a-c linear amplifier of the type comprising a final stage with a pair of cascaded output transistors operating in Class B or AB. The output transistors are driven in push-pull by way of an input stage and an intermediate stage, the input stage including a first or controlling transistor which drives during alternate half-cycles of the oscillations to be amplified so as to cut off a second or controlled transistor operating in Class A. One output transistor, i.e. the one closer to the high-voltage terminal of the supply, saturates when the second transistor is cut off.

BACKGROUND OF THE INVENTION

The output voltage generated by an amplifier of this type is substantially sinusoidal, except for the flattened peaks of its half-cycles resulting from the clipping effect due to saturation of the output transistors. The symmetry of the wave flanks, however, is disturbed specifically by the aforementioned second transistor to which the oscillations to be amplified are applied. As the first transistor overdrives the second transistor, the virtual capacitance at the input of the second transistor controlled thereby (formed between the base and the emitter of the latter transistor) is partly discharged so that a certain charging current must flow into that input capacitance at the end of the saturation interval before this second transistor will again conduct. This delays the desaturation of the one output transistor referred to above, whose operation is in phase with that of the first transistor in the input stage, leading to a distortion of the wave shape of the amplified output signal.

Such a distortion is particularly unwelcome when the cascaded output transistors are connected across an unbalanced d-c power supply; in that case, the harmonics generated by the distorted signal peak may strongly interfere with nearby electronic equipment, aside from entailing a substantial loss of power.

Since the delay in the recharging of the virtual input capacitance of the controlled transistor is determined by the circuit parameters and is practically independent of frequency, the resulting distortion becomes more significant at the higher frequencies of the operating range.

OBJECT OF THE INVENTION

The object of my present invention, accordingly, is to provide an improved amplifier of the above-described type in which this distortion of the output oscillation, especially at its high-voltage peak, is substantially avoided.

SUMMARY OF THE INVENTION

In accordance with my present invention I provide a normally nonconductive ancillary transistor connected to at least one of the amplifier stages for temporarily conducting upon cutoff of the controlled ("second") transistor and supplying same with a supplemental biasing voltage which maintains the controlled transistor at its conduction threshold throughout the time of saturation of the output transistor closer to the high-voltage terminal of the power supply.

When the controlled transistor has an emitter electrode connected to the low-voltage terminal and a collector electrode connected to the high-voltage terminal of the power supply (with inclusion of the usual resistors and/or constant-current generators in these connections) as well as a base electrode connected to the input stage of the amplifier, the ancillary transistor may have a base lead connected to a point of fixed biasing potential and an emitter lead connected to either the collector electrode or the base electrode of the controlled transistor, depending on whether the ancillary transistor is to function as a current generator or as a voltage generator. In the first instance the collector lead of the ancillary transistor may be connected to the base electrode of the controlled transistor or to a point of the input stage upstream of the controlling transistor; in the second instance this collector lead may be connected directly or indirectly to the high-voltage terminal of the power supply.

The connection between the collector electrode of the controlled transistor and the high-voltage supply terminal may further include a so-called bootstrap capacitor which boosts the potential of that electrode, during the saturation interval, to an absolute value higher than that of the high-voltage terminal itself, allowing the base of the ancillary transistor to be tied directly to the latter terminal.

BRIEF DESCRIPTION OF THE DRAWING

My invention will now be described in greater detail with reference to the accompanying drawing in which:

FIG. 1 is a graph showing a positive half-cycle of a-c oscillation in the output of a linear amplifier with and without the present improvement;

FIG. 2 is a circuit diagram of an amplifier embodying my invention;

FIGS. 3 and 4 are circuit diagrams similar to FIG. 2, showing two other embodiments;

FIG. 5 is a more detailed diagram of part of an amplifier such as that of FIG. 2 realized in integrated circuitry; and

FIG. 6 is a diagram similar to FIG. 5 but relating to a modified amplifier.

SPECIFIC DESCRIPTION

In FIG. 1 I have shown the upper half of a clipped sine wave, representing the output voltage V_(out) of a multistage linear amplifier, which rises sinusoidally from a mean level V_(m) to a flattened peak at a level V_(p) reached at a time t₁. This peak, not drawn to scale, theoretically lasts to an instant t₂ when the voltage curve begins to descend sinusoidally (dotted line) and symmetrically to its rising branch. In practice, however, the descent is retarded beyond instant t₂ and produces some undesired harmonics before rejoining the theoretical curve at instant t₃. This delay is the result of saturation phenomena, already discussed hereinabove, occurring mainly in one or more components of the amplifier ahead of its final stage.

In FIG. 2 I have shown an amplifier of the type here considered (generally similar to one shown in commonly owned U.S. Pat. No. 3,828,265) which comprises an input stage including a first NPN transistor Q₁, an intermediate stage with a second NPN transistor Q₂ connected to an NPN transistor Q₃ in a Darlington configuration, and a final stage with further NPN transistors Q₄ and Q₅ as well as associated pilot transistor Q₆ of NPN type and Q₇, Q₈ of PNP type. Output transistors Q₄ and Q₅ are connected between terminals 2 and 10 of a d-c power supply, terminal 2 carrying a positive voltage V_(S) whereas terminal 10 is grounded. The emitter of transistor Q₄ and the collector of transistor Q₅ are connected to a common central terminal 1 carrying the output voltage V_(out).

An oscillator labeled V_(in) symbolizes a source of a-c signals to be amplified, this oscillator being connected via a series capacitor C₁ between the base and the emitter of transistor Q₁ which is inserted between supply terminals 2 and 10 via a collector resistor R₄ and a constant-current generator U₂. A stack of diodes D₁ and another constant-current generator U₁ as well as a resistor R₁ form part of a biasing circuit for the base of transistor Q₁. A feedback path exists between output terminal 1 and the emitter of transistor Q₁, this path including a resistor R₅ and a capacitor C₂ in series.

The collector of transistor Q₁ is joined to the base of transistor Q₂ whose emitter is tied to the base of transistor Q₃ and is grounded through a resistor R₂, a similar resistor R₃ grounding the emitter of transistor Q₃. A "bootstrap" terminal 4 is coupled to output terminal 1 via a capacitor C_(B) and is connected to supply terminal 2 through a resistor R₇ ; terminal 4 feeds the collector of transistor Q₂ through a constant-current generator U₃ to which the collector of pilot transistor Q₃ is also connected by way of a stack of diodes D₂. The collectors of transistors Q₂ and Q₃ are further tied to the bases of pilot transistors Q₆ and Q₈, respectively, the latter transistor being linked with output transistor Q₅ in a composite configuration. Transistor Q₆, whose emitter is connected to output terminal 1, has its collector connected via a diode D₃ to bootstrap terminal 4 to which the emitter of transistor Q₇ is also connected; the latter transistor, whose base is tied to the collector of transistor Q₆, forms with diode D₃ a current mirror. A load R_(L) is inserted between output terminal 1 and ground in series with a capacitor C_(L).

In accordance with my present invention, an intermittently operative current generator G comprises an ancillary transistor Q_(O) (here of PNP type) having a base lead b connected to supply terminal 2, an emitter lead e connected to the collector of transistor Q₂ and thus via generator U₃ to the bootstrap terminal 4 carrying a voltage V_(B), and a collector lead c connected to the base of the latter transistor. At C_(c) I have indicated a virtual capacitance existing between that base and ground, this capacitance tending to be partly discharged when transistor Q₁ is saturated during positive half-cycles of voltage V_(in). In order to prevent such discharge, and thereby to avoid the delay discussed with reference to FIG. 1, the normally nonconductive transistor Q_(O) goes into conduction as soon as the output transistor Q₄ has gone into saturation and lead e becomes positive with respect to lead b with the cutoff of controlled transistor Q₂, thereby generating a collector current on lead c which recharges the virtual capacitor C_(c) to incipient conduction of transistor Q₂. When the transistor Q₄ thereafter desaturates, the ancillary transistor Q_(O) is cut off, at precisely the time when the transistor Q₁ begins to drive the transistor Q₂ into full conduction.

The amplifier shown in FIG. 3 differs somewhat from that of FIG. 2, corresponding components having been designated by the same reference characters. In this instance the oscillator V_(in) and its series capacitor C₁ are connected between ground and the base of a PNP transistor Q₉ whose emitter, like that of a similar transistor Q₁₀, is connected through a constant-current generator U₄ to supply terminal 2; the base of transistor Q₉ is biased by a voltage divider R₈, R₉, connected between supply terminals 2 and 10, via a resistor R₁₀. Transistor Q₉ lies in series with an NPN transistor Q₁₁ whose emitter is grounded via two resistors R₁₁, R₁₂ having a junction tied to the collector lead c of ancillary transistor Q_(O). The base of transistor Q₁₀ is connected to output terminal 1 through the feedback network R₅, C₂ which is here shunted by a series arm R₁₄, R₁₅ of a resistance network having a grounded shunt arm R₁₆. The collector of feedback transistor Q₁₀ is grounded through a diode D₄ in series with a resistor R₁₃ and is tied to the base of series transistor Q₁₁ forming therewith a current mirror. The base of controlled transistor Q₃ is connected to supply terminal 2 via a constant-current generator U₅ ; a similar constant-current generator U₆ feeds the emitter lead e of transistor Q_(O) through a resistor R₁₇ and the collector of transistor Q₃ through a diode D₂. PNP transistor Q₈ has its emitter connected to terminal 2 through an NPN transistor Q₁₂ in series therewith, the base of the latter transistor being connected to output terminal 1 through a stack of diodes D₇ and to supply terminal 2 through a constant-current generator U₇. Two further diodes D₅ and D₆ respectively connect the base of pilot transistor Q₇ to supply terminal 2 and to the emitter of transistor Q₆ which in turn is joined to output terminal 1 by way of a resistor R₁₈.

A fixed biasing potential for the base lead b of ancillary transistor Q_(O) is obtained in FIG. 3 from a voltage divider R₁₉, R₂₀ extending between the emitters of two NPN transistors Q₁₃ and Q₁₄, transistor Q₁₃ being connected as a diode between terminal 2 and the base of transistor Q₁₄ whose collector is tied to the same supply terminal and whose emitter is grounded through a constant-current generator U₈.

The biasing network including transistors Q₁₃, Q₁₄ and resistors R₁₉, R₂₀ (which may be considered part of the intermittently operable current generator G) is so dimensioned that transistor Q_(O) conducts only when its emitter is driven sufficiently positive upon saturation of output transistor Q₄, as in the preceding embodiment. In this instance, however, collector lead c does not directly charge the virtual capacitance C_(c) but supplies additional current to resistor R₁₂ so as to increase the voltage drop thereacross, thereby reducing the conductivity of transistor Q₁₁ and throttling the current flow through transistor Q₁ sufficiently to let generator U₅ bias the transistor Q₂ to its conduction threshold.

The system of FIG. 4 is substantially identical with that of FIG. 2, except that the current generator G including transistor Q_(O) has been replaced by a voltage generator H comprising not only this ancillary transistor but also a voltage divider maintaining the base lead b thereof at a predetermined potential between ground and supply voltage V_(S). This voltage divider is formed by a constant-current generator U_(O) and a stack of diodes D_(O) ; collector lead c is connected to supply terminal 2 in parallel with generator U_(O).

With emitter lead e tied to the base of controlled transistor Q₂, and thus also to the collector of controlling transistor Q₁, transistor Q_(O) conducts when the voltage on lead e drops below the conduction threshold of transistor Q₂ so as to maintain the base potential of the latter transistor at this threshold.

The integrated-circuit arrangement of FIG. 5 includes transistors T₄₄ -T₈₆ (some of them connected as diodes), Zener diodes Z₂ -Z₅, resistors R₃₀ -R₅₈ and capacitors C₄ -C₆. In addition to the aforedescribed terminals 1, 2, 4 and 10 there are also shown a jack 3 and terminals 5 and 7 connectable to external circuits for biasing or modulation purposes, along with a subsidiary terminal 9 resistively connected to ground in a nonillustrated part of the circuit. Transistors T₈₃ and T₈₄ play the part of output transistors Q₄ and Q₅ of FIGS. 2-4; controlling transistor Q₁ and controlled transistor Q₂ are represented by transistors T₆₁ and T₆₂, respectively. The ancillary transistor according to my invention (Q_(O) in FIGS. 2-4) is here represented by a transistor T₈₅ with its base (lead b) connected to supply terminal 2, its emitter (lead e) connected to bootstrap terminal 4 through resistor R₄₁ and transistor T₆₃, and its collector (lead c) connected to the base of controlled transistor T₆₂. Thus, the operation of the amplifier of FIG. 5 is analogous to that described with reference to FIG. 2. The bootstrap capacitor C_(B) of FIGS. 2 and 4 has not been illustrated.

The amplifier shown in FIG. 6, in which the aforedescribed terminals 1, 2, 4, 9 and 10 are supplemented by biasing or modulation terminals 6 and 8, comprises transistors T₂₀ -T₄₂ (again partly connected as diodes), resistors R₆₁ -R₇₆ and a capacitor C₇. The output transistors are here designated T₄₀ and T₄₁ ; transistors T₂₇ and T₂₈ respectively represent the controlling and controlled transistors. The ancillary transistor (Q_(O) in FIGS. 2-4) is shown at T₃₉ ; its base lead b is tied to supply terminal 2 while its emitter lead e is connected to bootstrap terminal 4 by way of transistor T₃₈ and resistors R₇₃, R₇₄ which together with resistor R₇₅ control the base voltage of output transistor T₄₀. Transistor T₃₉ conducts when that base voltage saturates the transistor T₄₀, coincidentally with the saturation of controlling transistor T₂₇ which cuts off the controlled transistor T₂₈ and thus drives the base potential of pilot transistor T₃₈ to its highest positive value. The conduction of transistor T₃₉ energizes its collector lead c which extends to ground through the emitter resistor R₆₄ of transistor T₂₄ ; thus, the corrective operation is analogous to that described with reference to transistor Q₁₁ and resistor R₁₂ of FIG. 3.

It will be apparent that the described and illustrated modes of intermittent energization of the ancillary transistor according to my invention can be modified and that the desired effect of its conduction upon the saturation of the final stage of the amplifier can also be realized with output connections of that transistor different from those specifically disclosed. Thus, for example, the emitter of ancillary transistor Q_(O) may be connected to either the input or the output of any transistor (such as final transistor T₄₀ in FIG. 6) upstream or downstream of controlled transistor Q₂. It is, of course, also possible to vary the base voltage rather than the emitter voltage of the ancillary transistor in order to render it conductive during the cut-off state of the second transistor only. 

I claim:
 1. In a linear a-c amplifier with a plurality of stages including an input stage, an intermediate stage and a final stage, a first transistor in said input stage receiving an a-c signal to be amplified, a second transistor operating in Class A under the control of said first transistor, and a pair of output transistors in said final stage driven in push-pull by said second transistor via said intermediate stage and connected in cascade across a d-c power supply for producing on a common central terminal an amplified replica of said a-c signal with alternate peaks clipped upon saturation of one of said output transistors coinciding with a cutoff of said second transistor during alternate half-cycles of said a-c signal,the combination therewith of a normally nonconductive ancillary transistor with an input connection to one of said stages for conducting only during the time of saturation of said one of said output transistors, said ancillary transistor being operatively coupled to said second transistor for biasing same to its conduction threshold prior to desaturation of said one of said output transistors.
 2. The combination defined in claim 1 wherein said power supply has a low-voltage terminal and a high-voltage terminal, said one of said output transistors being a third transistor inserted between said high-voltage and central terminals, the other of said output transistors being a fourth transistor inserted between said central and low-voltage terminals.
 3. The combination defined in claim 2 wherein said second transistor has an emitter electrode connected to said low-voltage terminal, a collector electrode connected to said high-voltage terminal via a constant-current generator, and a base electrode connected to said input stage, said ancillary transistor having a base lead connected to a point of fixed biasing potential and an emitter lead connected to an electrode of said second transistor.
 4. The combination defined in claim 3 wherein said emitter lead is connected to said collector electrode, said ancillary transistor further having a collector lead connected to said base electrode.
 5. The combination defined in claim 3 wherein said emitter lead is connected to said collector electrode, said ancillary transistor further having a collector lead connected to a point of said input stage upstream of said second transistor.
 6. The combination defined in claim 3 wherein said emitter lead is connected to said base electrode, said ancillary transistor further having a collector lead connected to said high-voltage terminal.
 7. The combination defined in claim 4 or 6 wherein the connection between said collector electrode and said high-voltage terminal includes a resistor whose junction with said constant-current generator is coupled to said central terminal via a capacitor driving said collector electrode to a potential higher in absolute value than that of said high-voltage terminal during said time of saturation.
 8. The combination defined in claim 7 wherein said point of fixed biasing potential is said high-voltage terminal. 